Measuring the level of an RF signal to be transmitted is important in transmitter devices that have a controllable amplifier section with variable gain, because accurately measuring the level of an amplified RF signal enables controlling the gain of the amplifier section so that it is at an optimal value at all times. In addition to the plain output power coming out of the last amplifier stage the levels of various reflected signals are of interest. For example an impedance mismatch somewhere between the power amplifier output and the antenna of a radio device causes reflections, which are unavoidable in the sense that even if perfect impedance matching at standalone conditions could be achieved, the position of the user's hand as well as the presence of conductive objects near the antenna change its impedance in practically unpredictable ways.
Multiradio is a widely employed overall designation of devices that may include a number of radio transceivers of different communication systems, like difference cellular radio systems, WLAN (Wireless Local Area Network), Bluetooth, and RFID (Radio Frequency Identifier) systems. The frequency ranges of these systems may be very much different from each other, while the constant strive for miniaturization and performance optimization in portable devices calls for combining as much of the transceiver functions, including RF power measurement to shared circuit elements to the largest possible extent.
A conventional way of measuring RF signal levels involves using a directional coupler, an example of which is shown in FIG. 1. The integrated RF circuitry comprises an active part 101 and a passive part 102. Within the active part 101 a radio frequency signal to be transmitted is conducted to a phasing part 103, which produces two versions of the radio frequency signal: a direct phase signal and another having a 90 degrees phase difference. These are separately coupled to the inputs of two parallel controllable amplifiers 104 and 105, which thus produce amplified in-phase and quadrature-phase versions of the radio frequency signal. The differently phased versions are combined in a so-called 3 dB hybrid 106, one output of which is terminated with a terminating impedance 107 while the other is coupled through a low-pass filter 108 and a directional coupler 109 to a load 110. The 3 dB hybrid 106, the terminating impedance 107, the low-pass filter 108 and the directional coupler 109 are located within the passive part 102.
In order to utilise the signal level information available at the directional coupler 109 there are detectors 111 and 112, which are located within the active part 101 and produce indications about the levels of the signals passing through the directional coupler 109, most importantly the output power level of the two-stage amplifier section. The location of the directional coupler 109 in the passive part 102 and the detectors 111 and 112 in the active part 101 necessitates additional connections between the passive and active parts, which is a drawback. To utilize the indications produced by the detectors 111 and 112 their outputs (designated as Vu and Vn in the drawing) are typically conducted to a control circuit, which in turn produces a control voltage Vc which controls the operation of the amplifiers through a controllable voltage source 113.
A major drawback of directional couplers is that they are difficult to implement in reasonable space into an integrated RF circuit. Trying to miniaturize a directional coupler leads usually to unacceptably high losses, poor directivity, too narrow bandwidth, and inaccuracy. It would be possible to compensate for inaccuracies with calibration, but only when impedances are constant. Calibrating a directional-coupler-based solution is not possible when unforeseeable changes occur in impedances, like in the case of the user's hand loading the antenna. Transformer type alternatives to basic directional couplers are known, with inductances and capacitances instead of transmission lines, but they share the same drawbacks related to large space requirements, and other difficulties in integrating with other circuit elements.
Attempts have been made to determine the power level of a signal to be transmitted also without a directional coupler, in some indirect way. Such attempts include measuring e.g. the DC power at the power amplifier, the bias current or the input signal level and trying to make deductions about the power levels at the output. The accuracy of such indirect measurements has been modest at its best.
FIG. 2 illustrates a prior art solution known from the publication US 20060205375 A1. A figuratively described signal generator 201 is adapted to generate a radio frequency signal and to have an output impedance Zo. The radio frequency signal generated by the signal generator 201 is conducted through a load, which may be for example an antenna and which has a load impedance Za. Between the signal generator 201 and the load there is a measurement circuit having an input 202 and an output 203. Coupled therebetween is a transmission line 204 that produces an exactly 90 degrees phase shift.
The radio frequency signal at the input 202 constitutes the first (in-phase) version of the radio frequency signal. Similarly the radio frequency signal at the output 203 constitutes the second (quadrature) version of the radio frequency signal. The first and second versions are taken to a phase shifter part which here consists of a pair of complementary phase shifters: the first 205 of the pair produces a −45 degrees phase shift, and the second 206 of the pair produces a +45 degrees phase shift. As a result, the two versions of the radio frequency signal have equal phases when they come to an adder 207.
The sum produced at the adder 207 does not depend on the reflection factor at all but only on the level of the original radio frequency signal generated by the signal generator 201. Thus an impedance mismatch between the signal generator 201 and the load does not affect the detection result Vs produced at the detector 208, which is indicative of the magnitude of the sum produced at the adder 207.
The drawback of the solution of FIG. 2 is that it is not easy to find a suitable location on the signal line where the exact 90 degrees phase shift would take place, and even if one is found, the required accuracy in the phase shift typically applies over a relatively narrow frequency range only. This problem becomes very prominent in multiradio devices, because the frequency range to be covered may be much wider than in conventional cellular phones where only the cellular channels need to be covered.